Memory circuit voltage regulator

ABSTRACT

As part of a memory array, a circuit is provided for altering the drive applied to an access transistor that regulates electrical communication within the memory array. In one embodiment, the circuit is used to alter the drive applied to a sense amp&#39;s voltage-pulling transistor, thereby allowing modification of the voltage-pulling rate for components of the sense amp. A sample of test data is written to the memory array and read several times at varying drive rates in order to determine the sense amp&#39;s ability to accommodate external circuitry. In another embodiment, the circuit is used to alter the drive applied to a bleeder device that regulates communication between the digit lines of the memory array and its cell plate. Slowing said communication allows defects within the memory array to have a more pronounced effect and hence increases the chances of finding such defects during testing. The circuit is configured to accept and apply a plurality of voltages, either through a contact pad or from a series of discrete voltage sources coupled to the circuit.

RELATED APPLICATION

This application is a divisional of application Ser. No. 08/855,555,filed May 13, 1997, and issued on Mar. 2, 1999, as U.S. Pat. No.5,877,993.

TECHNICAL FIELD

The present invention relates generally to semiconductor circuit devicesand, more specifically, to a circuit for changing the voltage applied toselective portions of a memory array. Such portions include digit linepairs as well as the gate of a transistor used to regulate senseamplifiers.

BACKGROUND OF THE INVENTION

In the operation of certain semiconductor circuit devices, pullup andpulldown sense amplifiers (sense amps) detect and amplify a small chargestored within a memory cell. In general, two complementary digit linesare attached to a pullup sense amp and a pull down sense amp. At thebeginning of a reading operation, both lines are at an equilibratevoltage Veq, which is generally between the potential of a voltagesource used to operate the semiconductor device (V_(CC)) and groundpotential (0 volts). While Veq is changeable either intentionally orinadvertently through a defect, Veq is ideally equal to V_(CC) /2 duringnon-test operations. This midpoint voltage is defined as DVC₂.

One of the digit lines is coupled to a memory cell. The reading processinvolves a discharge from the memory cell to the corresponding digitline, which creates a slight difference in voltage between the two digitlines. This difference is then amplified by the sense amps: the digitline with the slightly lower voltage has its voltage further decreasedby the pulldown sense amp, and the voltage of the other digit line isincreased by the pullup sense amp. Once the voltage difference has beenamplified, the digit lines can then be used to operate less sensitivecircuitry.

Between reading cycles, it is necessary to return the complementarydigit lines to Veq. This occurs during what is known as a prechargecycle, wherein equilibration transistors short the complementary digitlines together. Further, a signal having a potential of DVC₂ iscommunicated from a DVC₂ voltage generator to the shorted digit linesthrough a bleeder device.

Concerning the operation of the sense amps, it should be noted thatpulling down the voltage of a digit line involves coupling the line toground through a pulldown transistor. Because an entire row of digitline pairs often connects to the same puildown transistor through acommon node, the pulldown transistor will most likely have to drawcurrent from one line of each of several pairs. In doing so, there is arisk that the transistor will become saturated with current andtherefore become slower in pulling down the voltage of additional digitlines. This may lead to errors in reading, especially if an entire rowof memory cells is storing logic 1's except for one cell storing a logic0; for once the logic 0 is discharged, a slow pulldown may result in animproper reading of that logic 0 value.

One known way to solve this problem is to include an optional activearea in the gate of the pulldown transistor. The increased size of thegate raises the threshold at which the pulldown transistor becomessaturated. However, one of ordinary skill in the art will appreciatethat this solution requires a costly metal mask change. Further, anyattempt to speed up the slowed pulldown raises other problems inreading, as disclosed in U.S. Pat. No. 5,042,011, by Casper, et al. TheCasper '011 reference discloses that pulling down the common node tooquickly may result in capacitive coupling between the sources and drainsof the sense amp's transistors. During capacitive coupling, both digitlines in one sense amp are pulled down before the common node is pulleddown low enough to turn on one of the sense amp transistors. When thesense amp finally turns on, it shorts out the capacitive coupling,bouncing the digit lines and, in the process, creates line noise thatwill interfere with the ability to read the data properly.

Early saturation and capacitive coupling could be avoided if one knewthe margin--the difference in voltage between a logic 0 signal and alogic 1 signal--that the pulldown transistor was capable ofaccommodating. The only way to do so, as taught by the prior art, is toseparate the pulldown transistor with a laser and probe the gate.

As an alternative to determining the sense amp's margin, one couldsimply test the sense amp's ability to operate at the given sourcevoltage used in non-test operations. Prior art suggests entering aseries of test data patterns into memory. Logic 1's are written to thecells of each memory array, with the exception of one column of logic0's. As a result, each row contains only one cell storing a logic 0,thereby creating the most likely circumstance for an error in readingthe data. The data in the array is then read and checked for errors.Once the first group of test data has been processed, a second sample oftest data is entered with the logic 0's written to the next column. Thisprocess repeats until a logic 0 has been written to and read from everycell in any given row in the memory array. The results will indicate thepuildown transistor's ability to read data accurately. The problem withthis process, however, is that it is time consuming to enter multiplesamples of test data.

Thus, there is a need in the art for a quicker circuit and method fortesting the capabilities of a sense amp. Further benefit would bederived if this test could indicate the margin of the sense amp'spulldown transistor.

In addition to inadequate pulldown transistors, other problems, such asdefects arising during the processing of semiconductor devices, maycontribute to reading errors. Various techniques involving equilibrationof the complementary digit lines can be used during testing to detectthese problems. For example, occasionally a digit line willinadvertently have a short to ground. As a result, the potential of thatdigit line will leak towards 0 volts. To detect this problem, prior artteaches extending the time for the precharge cycle during a test mode.If the short has a low enough resistance, the short will overcome thecharging ability of the DVC₂ voltage generator, which remains coupled tothe digit lines, and Veq of the digit lines will decrease. Thus, alonger precharge cycle allows Veq to lower even further. As a result,line noise is more likely to register as a logic 0 discharge on thedigit line when in fact the storage cell contains a logic 1 and has notyet discharged. Alternatively, assuming that a logic 1 is properlydischarged and sensed, a reading error is still likely: Veq may be solow due to the short that the pullup sense amp may not be able to pullup the digit line's voltage in time to register as a logic 1 forpurposes of driving external circuitry. Increasing the likelihood oferror is desirable in the test mode, as it helps to identify errors thatwould affect non-test operations. Further, a reading error occunrrngafter this extended precharge cycle will indicate the nature of thedefect--in this case a short in at least one of the digit lines.However, this testing process can be time consuming. As an example, a 64meg DRAM having a 16 meg×4 configuration requires approximately 170seconds to carry out this test. It would be a benefit to the art to havea faster way to test for this problem.

A second problem that could be detected by altering the equilibrationrate of the digit lines involves a short between the cell plate and thedigit line. The typical technique for discovering this problem is toinitiate a long RAS (Row Address Strobe) low signal. During the low RAS,the digit lines are not equilibrated. Rather, they are charged to theircomplementary voltage levels. Ideally, once the low RAS ends and thelines are shorted, both digit lines should approach a Veq level of DVC₂.However, a short between one of the digit lines and the cell plate willallow the DVC₂ generator 68 to change that digit line's voltage duringthe RAS low period. Thus, once the lines are shorted, their respectivevoltages will meet at a different Veq level. This will affect the marginbetween Veq and the voltage corresponding to one of the logic values andthereby increase the likelihood of a reading error. Eventually, thesignal from the DVC₂ voltage generator will restore the properequilibrate voltage once the RAS low signal ends. Nevertheless, forpurposes of detecting this problem before non-test operations begin, itwould be desirable to slow the restoration of the proper Veq level.

A third example concerns a defect that could exist within the memorycell's storage capacitor, such as a defect in a nitride layer acting asa dielectric between the memory cell's conductive plates. Such a defectcould cause a short within the storage capacitor. Because the storagecapacitors are coupled to the DVC₂ voltage generator, a defectivecapacitor "storing" a 0 volt charge, representing a logic 0, will slowlycharge to the DVC₂ level. The closer the storage capacitor approaches aDVC₂ charge, the more likely that a logic 1 value may be misread duringthe next reading. One way to detect this problem in the prior art is toinitiate a static refresh pause, wherein the memory cell's accesstransistor remains deactivated for a longer time than usual--generally100 milliseconds. As a result, the capacitor, which should be storing alogic 0, has a longer time to charge to a higher voltage, thereby makingan error in the next reading cycle more likely.

Once again, a speedier test is desired. The defect might be detectedearlier if the problem were exacerbated to the point where the leakedcharge for the stored logic 0 exceeded the equilibrate charge of thedigit lines. As a result, a logic 1 would be read from the cell eventhough it was known that a logic 0 had been written. One could speed upthe leakage into the storage capacitor by forcing DVC₂ to a highervoltage. However, the equilibrate voltage of the digit lines would alsoincrease accordingly and remain higher than the voltage of the charge inthe storage capacitor. Thus, forcing DVC₂ would not appreciably increasethe ability to detect an error unless the equilibration of the digitlines could be slowed. The only way to do this in the prior art isthrough the use of a costly metal option to change the gate voltage ofthe bleeder device.

SUMMARY OF THE INVENTION

Given the need for regulating the drive of a sense amp, as well as theneed for regulating the equilibration signal from a DVC₂ voltagegenerator, a test circuit is provided for varying the voltage of asignal used to drive a connection device that allows electricalcommunication within a semiconductor circuit. One preferred circuitembodiment includes a contact pad for carrying a range of test voltagesignals to said connection device. In another preferred circuitembodiment, a regulator circuit enables a series of discrete voltages todrive the connection device.

In one set of applications involving the regulation of a sense amp, theconnection device comprises a sense amp's voltage pulling transistor.Any circuit embodiment covered by the present invention can be used totest drive the transistor. In a preferred method of use, a test datapattern is entered and the data is read several times, with a differentvoltage driving the sense amp's pulldown transistor each time. Oneadvantage of this preferred method is thal it reduces the need forentering several elaborate test data patterns and, therefore, allows forquicker testing of memory arrays. A second advantage is that theembodied method and devices allow a determination of the lowest supplyvoltage that can be used during normal operation without errors inreading data. Yet another advantage is the ability to determine thehighest supply voltage, and therefore the fastest reading speed, thatcan be used during normal operations without causing capacitivecoupling. In doing so, the preferred circuit embodiments and methodincrease the sense amp's ability to distinguish between a logic 0voltage and a logic 1 voltage without physically altering the sense amp.Further, in the process of determining the lowest and highest voltagesat which the sense arnp is capable of functioning, the preferredembodiments and method also provide a way to ascertain the marginwithout dissecting components of the sense amp.

Concerning the specific errors that may be detected in relation toequilibrating the digit lines, the connection device comprises anisolation bleeder device coupled between the DVC₂ voltage generator anda digit line pair. The circuit embodiments provide a test mode apparatusfor driving the bleeder device in order to slow or quicken theequilibration of the digit line pair. Applying these embodimentsprovides the advantage of a quicker detection of defects such as a shortfrom a digit line to ground, a short from a digit line to a cell plate,and a short within the storage capacitor of a memory cell. Theembodiments also provide an alternative advantage of overcoming theinfluence of these defects during non-test modes.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 depicts a row of n-channel pulldown sense amps with associated D,D*, and WL lines; a pullup sense amp; and a series of memory cells, asfound in the prior art. FIG. 1 also shows a digit line equilibrationcircuit as found in the prior art.

FIG. 2 is a graph indicating the voltage of the conductive paths D andD* over time in the event that a memory cell storing a logic 0discharges to D. FIG. 2 also demonstrates the resulting amplification ofthe difference in voltage.

FIG. 3 is a graph demonstrating the relationship between drive current(I_(DV)) and the gate-source voltage of a pulldown transistor (V_(GS))at various levels of voltage applied to the gate (V_(GATE)).

FIG. 4 details one exemplary circuit embodiment in accordance with thepresent invention as used with a sense amp.

FIG. 5 illustrates a second exemplary circuit embodiment in accordancewith the present invention as used with a sense amp.

FIG. 6 shows a third exemplary circuit embodiment in accordance with thepresent invention as used with a sense amp.

FIG. 7a is a schematic of a portion of a memory array depicting anembodiment of the current invention as used in the digit line/cell plateregion of a memory array. FIG. 7a further depicts a first type ofpossible defect within said memory array.

FIG. 7b is a graph illustrating the effect of the first defect and afirst embodied method of the current invention.

FIG. 7c is another graph illustrating the effect of the first defect andthe first embodied method of the current invention.

FIG. 8a depicts a cross-section of a portion of a memory array includinga second type of defect.

FIG. 8b demonstrates the effect on a memory array of the second type ofdefect as well as the effect of a second embodied method of the currentinvention.

FIG. 8c further demonstrates the effect on a memory array of the secondtype of defect as well as the effect of a third embodied method of thecurrent invention.

FIG. 8d depicts the effect of a fourth embodied method of the currentinvention as it relates to the second type of defect.

FIG. 9a is a schematic of a portion of a memory array depicting a thirdtype of defect in said memory array.

FIG. 9b is a graph indicating the effect of the third type of defect.

FIG. 9c is a graph illustrating a method in the prior art for detectingthe third type of defect.

FIG. 9d is a graph illustrating the effect of a fifth embodied method ofthe current invention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

FIG. 1 illustrates the general configuration of sense amps in a memoryarray. A pulldown sense amp 20 includes cross coupled n-channeltransistors Q1 and Q2, as well as a pulldown transistor Q3, which is ann-channel transistor driven by a signal designated as LENSA. Theseelements play a part in sensing and amplifying a voltage differencebetween D and D* caused by shorting a memory cell 22 to D by way ofaccess transistor Q4. The sources of Q1 and Q2 are connected to a commonpulldown node 24, and the gate of each is connected to the other'sdrain. The gate of Q1 also connects to the line D*, whereas the gate ofQ2 connects to the line D.

As discussed above, each line D and its corresponding line D* areinitially at the same voltage DVC₂. For purposes of explanation, DVC₂ isassumed to be 1.65 volts, or one half of the source voltage V_(CC),which is 3.3 volts. Lines D and D* connect to opposite sides of eachsense amp 20. Common pulldown nodes 24 found in the sense amp arrayswill also be at DVC₂. A signal sent through the path WL will cause astorage capacitor 150 of particular memory cell 22 to discharge to aline D, thereby slightly changing D's voltage while the voltage of D*remains at DVC₂. Again, for purposes of explanation, a memory celldischarge will be assumed to cause a 0.2 volt difference in D. Thepulldown sense amp 20 will then turn on when the common pulldown node 24is one transistor threshold voltage below D or D*, whichever is highest.For instance, if a memory cell 22 is storing a logic 1, a discharge to Dwill increase D's voltage to 1.85 volts. As a result, the puildown senseamp transistor gated by D (Q2) turns on faster than the one gated by D*(Q1). With transistor Q2 on, D*'s voltage is pulled down from 1.65 voltstowards ground as the common pulldown node 24 is pulled down as well.Further, the lowering voltage of D* serves to turn on the pullup senseamp transistor gated by D* (Q14) before the other pullup sense amptransistor turns on. The voltage supply V_(CC) then charges line D.

On the other hand, if the memory cell 22 had been storing a logic 0,then a discharge to D would slightly lower D's voltage to 1.45 volts.The pulldown sense amp transistor gated by D* (Q1) would turn on firstand D's voltage would be further decreased toward ground by the puildownsense amp, thereby allowing the pullup sense amp to increase D*'svoltage toward V_(CC). In this way, a small voltage difference between Dand D* is sensed and amplified. Once the voltage difference has beenamplified, D and D* can drive less sensitive circuitry not shown inFIG. 1. It should be noted that, if a logic 0 is transmitted to D, thenthe pulldown sense amp need only pull down D from 1.45 volts. If a logic1 is transmitted to D, then the pulldown sense amp must pull D* from thehigher DVC₂ level--1.65 volts.

Therefore, if many logic 1's in a memory array row are read, the extravoltage that must be pulled contributes to saturating the pulldowntransistor Q3 with drive current, thereby slowing any further pulldown.The problem created by slow pulldown is illustrated in FIG. 2, whereslope X denotes the initial discharge to D from a memory cell 22 storinga logic 0. FIG. 2 further illustrates the amplification of thedifference in voltage between D and D*. Slope Y denotes the timerequired for D to drop in voltage given a situation where a row of cellscontains a roughly equal number of logic 1's and logic 0's. Should therebe many Logic 1's read amongst a single Logic 0, then the outcomechanges: as the logic 0 is read, the pulldown transistor Q3, havingapproached saturation, takes much longer to pull down D's voltage. Thisresult is illustrated by slope Z. Other circuitry elements (not shown)that are driven by D may read D before its transition to a lower voltagehas been completed. As a result, a logic 0 value may be misread as alogic 1.

As illustrated in FIG. 3, increasing the voltage to the gate of thepulldown transistor allows the transistor to puildown more currentbefore saturation. One preferred embodiment of the current inventionthat uses this principal is detailed in FIG. 4, where the pulldowntransistor Q3 is driven by a test circuit 26 through an inverter 27. Inthis embodiment, the inverter 27 comprises a p-channel transistor Q6 andan n-channel transistor Q8. The coupled gates of inverter transistors Q6and Q8 form an input node 28 for receiving a signal ENSA*, which may beV_(CC), ground, or a signal from another driver. The coupled drains ofthe inverter transistors Q6 and Q8 output the LENSA signal that drivesthe pulidown transistor Q3. The source of Q8 is coupled to ground. Thesource of Q6 is coupled to a source node 30 that branches into a firstconducting path 32 and a second conducting path 34. The first conductingpath 32 is coupled to an n-channel transistor Q1O, which has a channelwidth-to-length ratio of around 500/2. The drain of transistor Q10 iscoupled to a contact pad 36. It should be understood that the term"contact pad" includes any conductive surface configured to permitelectrical communication with a circuit or a node. The gate oftransistor Q10 is coupled to an inverter 60 through another n-channeltransistor Q36. Together, inverter 60 and transistor Q36 comprise alatch device, and both are coupled to V_(CCP). Further, inverter 60receives a TEST* signal as an input. In addition, the gate of transistorQ1O is also coupled to a feedback capacitor 62. This feedback capacitor62 comprises an n-channel transistor having a size of approximately100/100, wherein the drain and source are shorted and coupled to thefirst conductive path 32. The second conducting path 34 is coupled to ap-channel transistor Q12, driven by a signal TEST, which is understoodto be the complement of TEST*. The transistor Q12 is also coupled toV_(CC), although no voltage source is considered to be a part of theinvention.

During testing, TEST* transmits a low voltage signal which is receivedby the inverter 60. In response, the inverter 60 initiates a V_(CCP)signal, sending it through transistor Q36 which outputs the V_(CCP)signal to the gate of transistor Q10, thereby switching on Q10. Thefeedback capacitor 62 serves to maintain and replenish this V_(CCP)signal in the event of leakage. Capacitive coupling between the gate anddrain of transistor Q10 allows Q10 to carry signals having a range ofvoltages for modifying the drive of the pulldown transistor Q3.Simultaneously, the TEST signal, applying a high voltage to transistorQ12, isolates V_(CC). A test data pattern is entered into the memorycells 22 and read with varying voltages driving the pulldown transistorQ3. The data read at various alternate voltages sent through bond pad 36can be compared with the data as originally written. This series ofreadings indicates the range of voltages through which the pulldowntransistor Q3 is capable of allowing accurate data readings. Oncetesting has ended, TEST* sends a high voltage signal and TEST becomeslow, thereby isolating the bond pad and allowing the V_(CC) signal totransmit to the pulldown transistor Q3.

The embodiment illustrated in FIG. 5 is a package part of thesemiconductor circuit device and receives a plurality of voltage sourceswith different magnitudes. The test circuit 26 allows selection amongthese sources for driving the gate of the pulldown transistor Q3. Theinverter 27 is the same as in FIG. 4. In this exemplary embodiment,however, source node 30 is coupled to three discrete voltage sources.First, source node 30 is coupled to V_(CCP) through a p-channeltransistor Q20 that is driven by a low signal A*. Source node 30 is alsocoupled to DVC₂ through another p-channel transistor Q22 that is drivenby a low signal B*. Finally, source node 30 is coupled to V_(CC) by wayof a p-channel transistor Q24. This p-channel transistor Q24 is gated bythe output of a logic unit, such as a NAND gate 46, which will drivetransistor Q24 in response to receiving a high signal A as a first inputand a high signal B as a second input. Given the input vector scheme ofthis embodiment, one of the transistors Q20, Q22, or Q24 will beoperable to the exclusion of the other two.

Thus, a low signal A* will drive the p-channel transistor Q20, therebyallowing V_(CCP) to drive the puildown transistor Q3. Simultaneously,signal B will be high, turning off p-channel transistor Q22. Further,the NAND gate output will also be high and turn off p-channel transistorQ24. If, on the other hand, signal B is low and signal A is high, thenonly p-channel transistor Q22 will be on, allowing DVC₂ to transmit tothe pulldown transistor Q3. Only when both signals A and B are high doesthe NAND gate 46 output a low signal and allow V_(CC) drive the pulldowntransistor Q3. The data read at these three voltage levels can then becompared with the data as originally written. It should be noted thatthis configuration does not require the die space needed for the contactpad 36.

Another embodiment concerns varying the voltage applied to a pullupsense amp 40. As seen in FIG. 1, the pullup sense amp 40 includes crosscoupled p-channel transistors Q14 and Q16 as well as a pullup transistorQ18. As one of ordinary skill in the art understands, there is generallya pullup sense amp 40 corresponding to every pulldown sense amp.Nevertheless, for purposes of clarity, only one pullup sense amp 40 isshown. The sources of Q14 and Q16 are connected to a common pullup node42, and the gate of each is connected to the other's drain. Further, thegate of Q14 connects to line D*, and the gate of Q16 connects to line D.Common pullup node 42 is coupled with pullup transistor Q18, which isanother p-channel transistor. Pullup transistor Q18 is also coupled tothe voltage source V_(CC). The pullup transistor Q18 is driven by asignal LEPSA*. FIG. 6 illustrates that the voltage driving pulluptransistor Q18 may also be varied through the use of a test circuit 26analogous to that used with the pulldown transistor Q3 in FIG. 5. FIG. 6depicts an inverter 27 comprising a p-channel transistor Q26 and ann-channel transistor Q28. The coupled gates of inverter transistors Q26and Q28 form an input pathway 48 for a control signal designated EPSA.The coupled drains transmit the inverted output signal EPSA* which, inturn, is received by a prior art device 50 that outputs the LEPSA*signal used to drive the pullup transistor Q18. The source of Q26 iscoupled to V_(CC), whereas the source of Q28 is coupled to the testcircuit 26 which, in this embodiment, includes three conductive paths.The first path 52 leads to DVC₂ by way of an n-channel transistor Q30,which is driven by a signal C. The second path 54 is coupled to avoltage source V_(BB) through an n-channel transistor Q32, as driven bya signal D. The third path 56 leads to ground by way of n-channeltransistor Q34. The gate of n-channel transistor Q34 is coupled to theoutput of a NOR gate 58. The NOR gate 58 accepts signal C as a firstinput and signal D as a second input and will activate transistor Q34only when both signals are low. Further, this embodiment is configuredin a manner analogous to the embodiment in FIG. 5, in that signals C andD will never simultaneously activate their respective transistors Q30and Q32.

The three n-channel transistors Q30, Q32, and Q34 will turn on if ahigh, or logic 1, signal is transmitted to their respective gates. Aswith the embodiment shown in FIG. 5 for the pulldown sense amp, thesignals and transistors are configured to allow only selectivecommunication between one voltage source and the pullup transistor Q18.As a result, if signal C is high, it will latch the n-channel transistorQ30 and provide electrical communication between DVC₂ and the pulluptransistor Q18. At the same time, the low signal from D turns offn-channel transistor Q32. Under these circumstances, the signals C and Dalso result in a low signal output from the NOR gate 58, thereby turningoff n-channel transistor Q34. Thus, all of the other voltage sources areisolated. Similarly, if signal D is high, then only n-channel transistorQ32 is turned on and V_(BB) electrically communicates with pulluptransistor Q18. When both signals are low, the NOR gate 58 outputs ahigh signal, thereby grounding the source of the n-channel invertertransistor Q28. This embodiment has benefits similar to the embodimentin FIG. 5.

Returning to FIG. 1, a prior art equilibration circuit can be seen aspart of the memory device. For purposes of explaining the followingembodiments of this invention, V_(CC) is now presumed to be 5 volts. Atransistor Q101 is coupled between digit line D and its complementarydigit line D*. The transistor is driven by an equilibration signal EQ.It should be noted that the signal EQ results from a logic function andis distinguishable from the equilibrate voltage Veq, which representsthe common mid-range voltage level of the complementary digit linesbefore a reading operation.

The signal EQ also drives two additional transistors Q102 and Q103,which are connected together in series at a node 120. These connectedtransistors Q102 and Q103 are also coupled between lines D and D*.Moreover, node 120 is coupled to a cell plate 64 and a DVC₂ voltagegenerator 68 through a bleeder device 122. The DVC₂ voltage generator 68transmits a cell plate signal CP of voltage DVC₂ to the node 120. Forpurposes of explaining the following embodiments of this invention, DVC₂is now 2.5 volts. The bleeder device 122 is driven by a signal ofvoltage V_(CCP), wherein V_(CCP) results from having pumped V_(CC) to aneven higher potential.

At the beginning of a precharge cycle, digit line D and itscomplementary digit line D* are at different voltages as a result of adischarge of the memory cell 22 during the reading cycle. One line willhave a charge equal to the V_(CC) value of 5 volts, while the other linewill have a 0 volt charge. The equilibrate signal EQ is then sent,activating transistor Q101, which shorts D and D* together. Moreover,the signal EQ activates transistors Q102 and Q103, which not onlyprovide another short between D and D* but also allow the CP signal tobe communicated to those lines. As a result, the lines D and D*equilibrate, both gaining a charge of potential DVC₂ (2.5 volts), whichis the desired equilibrate voltage Veq in this-example. Once the linesare equilibrated, they are ready for further testing.

For various reasons, a particular portion of the memory array may bedefective. Hopefully, testing processes will identify those defects. Asdiscussed above and illustrated in FIG. 7a, a first defect 124 that mayexist is a short to ground of the digit line D. FIG. 7b illustrates theeffect of the first defect 124. During the precharge cycle, the CPsignal is trying to charge the digit lines D and D* to the 2.5 volt DVC₂level and maintain that level. However, if the resistance of the shortis not too great, the first defect 124 may cause the digit lines todischarge toward ground faster than CP can charge them to 2.5 volts. Asa result, once the precharge process has ended at time t₁, the digitlines may be equilibrated at a potential lower than 2.5 volts, such as1.7 volts. Having a Veq at a level other than DVC₂ makes the memoryarray susceptible to reading errors. For example, in the presentsituation illustrated in FIG. 7b, where Veq is too low, line noise on Doccurring at time t₂ is more likely to register as a logic 0 dischargewhen in fact the storage cell 150 contains a logic 1 and has not yetdischarged. Alternatively, assuming that a logic 1 is properlydischarged and sensed at time t₂ ', a reading error is still likely: asseen in FIG. 7c, Veq may be so low due to the short that the pullupsense amp may not be able to sufficiently pull up the digit line'svoltage by the time t₃, when external circuitry accesses line D. Inorder to find such a reading error, prior art requires an extendedprecharge time, up to time t₁, in order to allow the discharge from thefirst defect 124 to overtake the charge from CP.

The current invention, however, provides an alternative to requiring along precharge time. FIG. 7a illustrates that the V_(CCP) signal drivingthe bleeder device has been replaced with the test circuit 26 thatapplies a different voltage V_(REG) to regulate the bleeder device. Inthe case of the first defect 124, the test circuit 26 transmits a signalhaving a voltage lower than V_(CCP) to drive the bleeder device 122.This causes a slower charge rate and allows the discharge from the firstdefect 124 to quickly overtake the charging from CP, as seen by thedashed lines in FIGS. 7b and 7c. With the resulting increased disparitybetween the charge rate and the discharge rate, the precharge periodneed only endure until time t₁ ' in order to increase the likelihood ofdetecting an error.

The design of test circuit 26 can be the same as those used in FIGS. 4and 5, wherein a source node 30 has access to at least one test voltage,either through a bond pad 36 or from a discrete voltage source. In thisapplication, however, the source node 30 is coupled to the bleederdevice 122. Furthermore, V_(CCP) is the voltage used in non-testoperations to drive the bleeder device, and V_(CC) and DVC₂ are used toslow the charge rate. It should be further understood that the number ofvoltage options could be increased. Alternatively, the number of voltageoptions could be decreased to offer only one test voltage and onenon-test voltage.

These circuit embodiments, as well as others falling under the scope ofthe invention, have uses in detecting other defects. FIG. 8a illustratesanother defect 136 that might occur within a memory array. Thecross-sectional view in FIG. 8a shows the cell plate 138 coupled to afirst n-region 140 of access transistor Q4. Ideally, the only way forthe DVC₂ voltage generator 68 to charge the digit line D through thecell plate 138 is to drive the gate 142 of transistor Q4 so that thecharge may pass from the first n-region 140 to a second n-region 144.From there, the charge travels through a tungsten plug 146, which servesas a contact between the second n-region 144 and the digit line D.Occasionally, however, a second defect 136 in the memory array may occurin the form of a short between the cell plate 138 and the tungsten plug146. As discussed above, a long RAS low signal is used to detect thissecond defect 136. Assuming line D is charged to 0 volts, FIG. 8b showsthat the long RAS signal allows line D to be charged to a highervoltage. Thus, when the low RAS signal ends at time t₁ and the digitlines are shorted to begin equilibration, the digit lines will no longerhave an initial tendency to reach an average potential between 5 and 0volts (2.5 volts). Rather, because line D is now higher than 0 volts,the shorted lines will settle at a higher midpoint, such as 3.5 volts.At this point, the margin between the new equilibrate voltage and thevoltage representing a logic 1 has decreased. Thus, an erroneous readingis more likely, as discussed above.

Conversely, if line D is initially charged to V_(CC) (FIG. 8c), theshort to the cell plate will cause D's voltage to lower during a longRAS low period. The resulting equilibrate voltage of lines D and D*could be lower than the preferred 2.5 volts. The lower equilibrate wouldagain make an error in reading more likely. In either case, the CPsignal will restore the equilibrate voltage to 2.5 volts by time t₂.However, by decreasing the drive to the bleeder device 122, any of theembodiments of the current invention will serve to slow down therestoration of Veq to DVC₂. With restoration time extended to time t₂ ',any circuit embodiment of the current invention increases the likelihoodof detecting errors that would suggest the existence of the seconddefect 136. Alternatively, FIG. 8d shows that a circuit embodiment ofthe current invention could be used during a non-test mode to compensatefor the second defect 136 by driving the isolation device 122 at ahigher-than-normal level. As discussed above, the bleeder device 122 isnormally driven at V_(CCP), a voltage level representing one or twoV_(t) 's above V_(CC). The potential V_(t), in turn, is the thresholdvoltage of the bleeder device 122. A further increase in the potentialof V_(CCP) would allow the bleeder device 122 to quickly restore Veq to2.5 volts by time t₂ ". The shorter restoration period reduces thechances of an erroneous reading.

FIG. 9a demonstrates yet another instance wherein the current inventioncould shorten test time. This instance concerns a third defect 148comprising a short that may be caused by a nitride defect within thestorage capacitor 150 of a memory cell 22. It should also be noted thatone of the plates of the storage capacitor 150 is in fact the cell plate138 on 64 and is therefore connected to the DVC₂ generator. Given thisthird defect 148, FIG. 9b indicates that the CP signal, having apotential of DVC₂, will charge the storage capacitor 150 toward thatpotential even though a logic 0 has been written to that cell for testpurposes. During a static refresh pause, the word line WL leading to thememory cell 22 will continuously transmit a low signal, which turns offaccess transistor Q4 of the memory cell 22 and allows the storagecapacitor 150 to take on a greater charge. With the stored charge havinga higher voltage, such as 2 volts, it is more likely that the logic 0will be misread at line D as a logic 1. In order to speed up the leakageinto the storage capacitor 150, DVC₂ is forced to a voltage higher thanthe normal 2.5 volts. Unfortunately, this would not result in muchbenefit under the prior art, as demonstrated by FIG. 9c: because the CPsignal has a voltage of DVC₂ and is in communication with D and D*during the static refresh pause, the CP signal would also charge lines Dand D* to a higher voltage. With the circuit embodiments of the presentinvention, however, a lower voltage could be used to drive the bleederdevice 122 and thereby slow the charging of the digit lines, asillustrated in FIG. 9d. Thus, while D and D* are regulated tosubstantially remain at 2.5 volts despite the forced DVC₂ voltage, thestorage capacitor may be quickly charged to a higher potential, such as2.7 volts, which exceeds the equilibrate voltage and makes it verylikely that a logic 1 will be mistakenly recognized.

One of ordinary skill can appreciate that, although specific embodimentsof this invention have been described for purposes of illustration,various modifications can be made without departing from the spirit andscope of the invention. Concerning the invention as used with a senseamp, for example, a test circuit for the pullup sense amp could beconfigured to transmit an entire range of voltages through a contactpad, as done with the pulldown sense amp depicted in FIG. 4. Inaddition, the test circuit 26 in FIG. 6 could be used with a pulldownsense amp. Conversely, the test circuit 26 in FIG. 5 could be used witha pullup sense amp. Moreover, both of these test circuits could becoupled to the same inverter and used to test drive either type of senseamp.

Further, regarding the embodiments use with a cell plate, it should benoted that the embodiments may be applied for other testing. Any circuitembodiment, for instance, may be used during the precharge cyclediscussed above in order to detect a short between a row line and acolumn line. Moreover, a circuit embodiment of the current inventioncould also be used during a non-test mode to overcome other defects inaddition to the short between a digit line and cell plate, as describedabove.

It should also be noted that, given a particular voltage source used inan embodiment, that source can be independent of V_(CC) rather than amere alteration of V_(CC), such as V_(CCP) or DVC₂. Accordingly, theinvention is not limited except as stated in the claims.

what is claimed is:
 1. A test mode drive-modifying device for a memoryarray having at least one sense amp coupled to a voltage-pullingmechanism, comprising:a test voltage alteration mechanism, wherein:saidtest voltage alteration mechanism is coupled to said voltage-pullingmechanism; and said test voltage alteration mechanism is configured toreceive a plurality of voltages.
 2. The test mode drive-modifying devicein claim 1, wherein said sense amp is a pulldown sense amp.
 3. The testmode drive-modifying device in claim 2, wherein said voltage-pullingmechanism is a transistor.
 4. The test mode drive-modifying device inclaim 3, wherein said voltage-pulling mechanism is an n-channeltransistor.
 5. A test mode driver circuit for a voltage-pullingtransistor of a sense amp, comprising:a conductive path coupled to agate of said voltage-pulling transistor; and an apparatus configured toreceive a range of voltages coupled to said conductive path.
 6. The testmode driver circuit in claim 5, wherein said apparatus is a contact pad.7. A voltage-pulling circuit for a sense amp, comprising:a mainconductive path coupled to said sense amp; and a plurality of secondaryconductive paths selectively coupled to said main conductive path,wherein each of said plurality of secondary conductive paths isconfigured to receive a generally discrete voltage.
 8. Thevoltage-pulling circuit in claim 7, wherein said sense amp is a pullupsense amp.
 9. A driving circuit for a voltage-pulling transistor of asense amp, comprising:a selective coupling apparatus coupled to andelectrically interposed between:a test conduit configured to accept aplurality of voltage signals; and an output node configured to couple toa gate of said voltage-pulling transistor; wherein said selectivecoupling apparatus has a first mode of operation and is configured toreceive a signal that has a first value during said first mode ofoperation, and wherein said selective coupling apparatus is furtherconfigured to allow electrical communication between said test conduitand said output node in response to said first value.
 10. The circuit inclaim 9, further comprising a main conduit coupled to said selectivecoupling apparatus and configured to receive a voltage source, andwherein:said selective coupling apparatus has a second mode ofoperation; said signal has a second value during said second mode ofoperation; and said selective coupling apparatus is configured to allowelectrical communication between said main conduit and said output nodein response to said second value.
 11. The circuit in claim 10, whereinsaid selective coupling apparatus is configured to prevent electricalcommunication between said test conduit and said output node in responseto said second value.
 12. The circuit in claim 11, wherein saidselective coupling apparatus is configured to prevent electricalcommunication between said main conduit and said output node in responseto said first value.
 13. The circuit in claim 12, wherein said secondvalue is inverse to said first value.